DigiSpeaker - The end of analog music

Great Resource

tylerjbrooks — Mon, 01 Mar 2010 14:47:39 -0800

Abdul Sharif over at Schippers and Crew is a great resource!
He located a couple local precision sheet metal shops for me today.
Thanks Abdul.

P7 - Power Supply Finished

tylerjbrooks — Sat, 20 Feb 2010 16:20:17 -0800

I have completed my experiments with the Prototype 7 Power Supply.
Here is the schematic.
I have left out the multiple output stages and concentrated on the +5V output. The trouble I was having before stemmed mostly from the transformer construction. Specifically, the leakage inductance was way too high for the size of the primary winding inductance. This was causing large spikes in the current through the sense resistor which would, in turn, tell the controller to turn off. By increasing the inductance and reducing the leakage inductance, I was able to get stable output regulation.

The trick to doing this experiment on a breadboard is to work hard at controlling the leakage inductance of the transformer. I tried several different winding techniques. Two things made a huge difference. First, a 'progressive' style of winding is much better than a 'straight' style. The difference is well explained in section 3.5.9 of Marty Brown's book (The Power Supply Cookbook). In short, say you have to do 90 turns to make a transformer. Furthermore, say the bobbin you have selected only has room for 30 windings. It is much better to wind three turns vertically (one on top of the other) than it is to wind and entire row, then overlap another row and finally overlap another row (to make 3 x 30 = 90 turns). The reason for this is that with straight winding, by the time the second row returns to overlay the first winding (so turn 60 is overlaying turn 1), the voltage difference in the wires at that point are large (assuming the voltage drop across the transformer is evenly distributed). This creates a capacitance between the two windings that increases the leakage inductance. Notice that except in the middle, just about every winding in a straight winding approach has this problem. Alternatively, if you just 'scrunch' all the windings on top of one another (as best you can) and progressively cross the bobbin such that turn N, turn N+1, turn N+2, ...etc are right next to one another, then the leakage is greatly reduced.

Secondly, splitting or doubling the winding between the inner core and the outer layer of the bobbin reduces leakage. Say, for instance, that you need to do 60 windings. By putting 30 as the inner winding and 30 as the outer winding and connecting the serially the leakage is reduced. Alternatively, you can put all 120 windings on the inner layer and then do another 120 windings on the outer layer and connect them in parallel. This last approach (parallel windings) seemed to work the best but required a lot more wire.

New Layout. Here is the new layout. It was suggested on the diyAudio site that I try to compact my design to reduce some of the noise. I had mixed success with this but it makes for a cleaner looking design. The parts that still have their 'legs', in general, are snubber part that I am constantly playing around with so I don't trim them down.
Vds and Ids. This is an example of Vds and Ids at medium load. Notice that I still have a lot of dv/dt trash but it isn't as bad as it was before. By increasing the inductance of the primary winding and reducing the leakage inductance, I was able to make a stable output.
Vds and Output. This is an example of Vds and the +5V otuput at medium load. I still have rather large spikes in the output but I believe I could get those under control with better construction (better xformer and a PCB).
Vds 'Bump'. I still get some unexplained behaviour. In this picture, notice that I get a 'blip' or 'bump' in the Vds just after a successful cycle. This is caused by a false trigger of the gate. That false trigger appears to happen when I get a 'ground bounce'. My ground and voltage rail coming from the rectifier section seem to get a rather large spike which messes up my 5 volt reference from the controller. Bouncing that reference (and the ground) will cause the voltage at pins 1 and 2 to cross momentarily which is what makes the gate fire. I don't know the cause of this but I felt it was time to move on to another design.

P7 - Power Supply Built

tylerjbrooks — Sat, 23 Jan 2010 01:34:12 -0800

I simplified the schematic in the previous post and built up the circuit on a breadboard.
Here is Schematic A.
I have also included the basic design information in the schematic.

UPDATE: I have redone the schematic again. I now bypass the error amplifier in the UC3845. I removed the +15VDC and +20VDC secondary to simplify testing. I have included snubbers on the primary side.
Here is Schematic C.

The Board. VAC comes in at the top left. The top row is the line filter and AC rectification. The second row has the controller, the power switch and the transformer output. The third row has the optoisolator and feedback control logic. I wound the transformer by hand. It has five separate windings (primary, bias, +5V, +15V, +20V). Each winding is insulated from the other (my first attempt was prettier but it shorted out during a current spike -- always insulate!). The transformer is 'gapped' with 4 pieces of electrical tape. The exact gap was a process of trial and error. I used an RLC circuit to measure the inductance. I had computed that I needed a 450uH inductor so I simply kept adding more gap (more tape) until I got the inductance to come down to 450uH. Of course, this added to the leakage inductance (~26uH).	Built PSU Schematic A - No Snubbers	Built PSU Schematic C - With Snubbers
Here is a picture of the MOSFET drain-to-source voltage (Vds) while regulating a 500mA load on the +5V line. The yellow trace is my Vds and the blue trace is the MOSFET gate voltage. If you know anything about these waveforms then you know this one is a little bit of a mess. What is happening is that during the on-time (when the blue trace is low), the current in the inductor is building and the current in the output (the secondary) is blocked by the secondary diode (the output capacitor holds the voltage). When the MOSFET turns off, the voltage on the primary reverses/spikes to nearly 600V!. This starts the secondary current which build for about 1uSec, then decays for about 4.5uSec. When that ends, the controller goes into discontiguous mode (runs out of current) and it decays to the input voltage over the remainder of the cycle. To fix this, I need to adjust the controller so that is uses more of the switch period (closer to contiguous mode) and add a primary snubber circuit to clip off the 600V spike.	Yellow = Vds Voltage Blue = Gate Schematic A - No Snubbers	Yellow = Vds Voltage Blue = Gate Schematic C - With Snubbers
Here is a picture of the MOSFET source current as measured through the sense resistor. The sense resistor is 0.25 Ohms so dividing any voltage reading by 0.25 will yeild the current. The current ramps from near zero at -6.72uSec (-6.72uSec, -16mV) up to 576mV at -0.28uSec (-0.28uSec, 576mV). During this time, the voltage across the inductor is 150V (nearly. There is a drop across the switch because this switch has a fairly high Rds ~= 2.2Ohms). The voltage across the inductor is equat to the inductance times the change in the current divided by the time (v = L * di/dt). Using this equation, it follows that the inductance is 431uH (=150 * 6.4uSec / 2.24A). This is close to the inductance that was measured using an RLC circuit (=467uH) (at least, within the measurement capabilities of my equipment). Notice the ugly spikes of current at the start and the end of the ramp. They go well over 4Amps! These have to be cleaned up. This inject a lot of noise into the output.	Yellow = Ids Blue = Gate Schematic A - No Snubbers	Yellow = Ids Blue = Gate Schematic C - With Snubbers
+5VDC Output. Both of these pictures were taken with Schematic C (snubbers included). The left picture does not include the output inductor (L2 - 2.2uH). The right picture does.	Yellow = Vds Blue = +5VDC Output Schematic C - With Snubbers - Without Output Inductor	Yellow = Vds Blue = +5VDC Output Schematic C - With Snubbers - With Output Inductor

P7 - Power Supply

tylerjbrooks — Mon, 11 Jan 2010 00:20:08 -0800

I am starting a new prototype. The new designation is 'Prototype VII' -- or 'p7' for short.

I thought I would try a 65W flyback switch mode power supply. The idea would be to put one of these into each DigiSpeaker. The flyback topology is compact, low part count and cheap while being relatively high in efficiency and performance.

Click here for the schematic.
Click here or here for the BOM. Q1 price is $22.35. Q1K price is $9.90. Both prices are without transformer, various connectors and so on.

The Design (procedure taken from the Power Supply Cookbook, Marty Brown):

Input	90VAC to 240VAC @ 50/60Hz
Output	Three Outputs: Output1: +20VDC @2.25A, Ripple: 100mVpp, Regulation: +-5% Output2: +15VDC @.333A, Ripple: 100mVpp, Regulation: +-10% Output3: +20VDC @3.00A, Ripple: 100mVpp, Regulation: +-10%
Total Output Power	(20V2.25A) + (15V.333A) + (5V3A) = 65W*
Total Input Power Flyback topologies tyically get 80% efficiency.	65W / 0.8 = 81.25W
DC Input	110VAC -> Vlow = 90 * 1.414 = 127VDC 110VAC -> Vhigh = 130 * 1.414 = 184VDC 220VAC -> Vlow = 185 * 1.414 = 262VDC 220VAC -> Vhigh = 240 * 1.414 = 340VDC
Average Input Current	Iin_high = 81.25W / 127VDC = 0.639A Iin_low = 81.25W / 340VDC = 0.239A
Peak Current	Ipeak = 5.5(65W) / 127VDC = 2.815A
Heat MOSFETs typically have 35% of the losses. Rectifiers typically have 60% of the losses.	Total Loss = 81.25W - 65W = 16.25W MOSFET Loss = 16.25W * 0.35 = 5.68W 20V Rectifier Loss = (45/65W) * 16.25 * 0.6 = 6.75W 15V Rectifier Loss = (5/65W) * 16.25 * 0.6 = 0.75W 5V Rectifier Loss = (15/65W) * 16.25 * 0.6 = 2.25W
Transformer	Primary Inductange = (127VDC * 0.5) / (2.815A * 50KHz) = 452uH Core Gap = (0.4 * PI * .452mH * 2.815A * 10*8) / (0.904 20002) = 0.044cm (or 17mils) Core Selection: Magnetics, Inc. 0F43007EC and 0F43007G044** Primary Turns = 1000 * (.452/100)0.5 ~= 67Turns** 20V Secondary Turns = (67 * (20 + 0.5) * (1 - 0.5)) / (127 * .5) ~= 11Turns 15V Secondary Turns = (15 + 0.9)(11Turns) / 20.5 ~= 9Turns 5V Secondary Turns = (5 + 0.9)(11Turns) / 20.5 ~= 3Turns
Output Filter	20V Reverse Voltage = 20V + (11T/67T)340VDC = 75.8V @ 2.25A -> MUR420* 15V Reverse Voltage = 15V + (9T/67T)340VDC = 60.7V @ 0.333A -> MUR120* 5V Reverse Voltage = 5V + (3T/67T)340VDC = 20.2V @ 3.0A -> MUR420* 20V Output Capacitor = (2.25A * 18uS) / 100mVpp = 405uF -> 2x 220uF @ 35V 15V Output Capacitor = (0.333A * 18uS) / 100mVpp = 60uF -> 1x 100uF @ 25V 5V Output Capacitor = (3.0A* 18uS) / 100mVpp = 270uF -> 1x 220uF @ 10V
Power MOSFET	Vdss > 340 + (67/3)(5+0.5) = 462V Ipeak < 3A Select: STMicro STP4NK50ZD
Feedback Regulation	Start by assuming a 1mA regulation current per volt. R1 = 5V/5mA = 1Kohm R2 = 5 - (2.5 + 1.4) / 6mA = 180ohm R3 = 2.5V / 1mA ~= 2.7Kohm Isense = 2.5V / 2.7Kohm = 0.926mA Spread the regulation between all three outputs. 20V and 15V get 40% of the regulation each. 5V get 20% of the regulation. R4 = (5V - 2.5V) / 0.2(0.926mA) = 13.5Kohm R5 = (15V - 2.5V) / 0.4(0.926mA) = 33.75Kohm R6 = (20V - 2.5V) / 0.4(0.926mA) = 47.25Kohm
Current Sense	Rcs = Vcs / Ipeak = 0.7V / 2.815A = 0.249ohms @ 2.4W
Feedback Loop	20V Pole = 1 / (2 * PI * (20/0.6) * 440uF) = 10.85Hz 15V Pole = 1 / (2 * PI * (15/0.1) * 100uF) = 10.61Hz 5V Pole = 1 / (2 * PI * (5/1.0) * 220uF) = 144Hz ADC = ((340 - 5)*2 3T) / (340 * 67T) = 14.78 GDC = 20log(14.78) = 23.4dB Gxo = 20log(10KHz/10.85Hz) - 23.4 = 35.89db or 62.31 So... C1 = 1 / (2 * PI * 13.5Kohm * 62 * 20KHz) = 9.5pF R2 = 13.5Kohms * 62 = 840Kohms C2 = 1 / 2 * PI * 10.85Hz * 840Kohms = 17.4nF

Prototype VI - Outdoors

tylerjbrooks — Thu, 07 Jan 2010 15:22:54 -0800

So, Jon and I have been curious about how much of our system response is due to the electronics/speakers and how much is due to the room. In the previous post, you can see that there are a number of spikes and nulls in the response. The room correction filter helps out a lot. However, we were wondering if we could put our system in an anechoic chamber and see the same (nearly same) response.

I posted a question on the Digital Room Correction mailing list and a user named Gregory Maxwell said that most of the response was due to the room. I wanted to put this to the test so I took prototype VI out on my deck for a test.

Bottom Line: Greg was right, most of the nulls and spikes in the response seem to be caused by the room.

Here are a couple pictures of my setup and a screen capture of the responses in Audacity.

Setup. I live on a hill above a lake. My deck is about 12 feet off the ground. From its corner, there is an unobstructed view to the lake. The closest possible reflector is more than 50 feet away and even at that, there are numerous bushes and other 'diffusers' to cancel echoes. I brought my lab computer, bench power supply and the amplifier (TAS5504+TAS5142) to the deck. These are the same parts I use in my lab for previous measurements. Unfortunately, my neighborhood is not very quiet. They are building a house down the block from me (hammers, power tools, ..etc). Cars drive by all the time. The landing pattern for the local airport goes over my neighborhood. Finally, the city seems to just generate a dull/low hum all the time. I have left some of the ambient noise on the end of the traces below so you can get an idea of its magnitude
Setup from Below. This is a picture from the yard below my deck. It gives you a good prospective on how free/clear the area is in front of the speaker.
Microphone. I attached the Behringer ECM8000 microphone to the end of a broom handle. I then asked my wife to hold it about 1 meter directly in front of the speaker while I conducted the experiment (sorry, no pictures of her doing this. We were both too busy.). Of course, this is not very precise but I figured it would be good enough to get a gross sense of the response outdoors so I could compare it with the indoors response. She tried her best, but she was not able to hold the microphone perfectly still. So, as she 'wobbled' it back and forth (about a 2 inch movement), I am sure it effected the response. Next time we get a sunny day (no rain), I will take the time to build some fixed mount for the microphone.
Response. The top trace is the logsweep. The second and third traces are typical recordings done in my lab. The fourth trace is a typical trace done outside. As you can see, many of the nulls and spikes are gone. The spike at 3000Hz (about 33.5 seconds) is our midrange to tweeter crossover. Our woofer to midrange crossover is at 200Hz (at 17.5 seconds) and is really not visible. The 300Hz null (at 20 seconds) that has always been present in the recordings I did in my lab is gone. This is strong proof that the 300Hz null is just a natural dead spot in my basement lab. O... notice that the trace is very noisy. Not much I can do about the noise in my neighbourhood.